Vector network analyzer applique for adaptive communications in wireless networks

ABSTRACT

A test signal generator at a transmitter station and a facsimile generator at a receiver station go through an acquisition and tracking process which aligns the two signals so that a logical processor can compute the frequency transfer function of the entire propagation path for use in an adaptive, concurrently sent communication signal. The frequency transfer function is conveyed back to the transmit end via a control channel permitting an adaptivity function at the transmit end to influence subsequent selection of communication parameters, among which are typically transmitted data rate, selection of modulation, selection of forward error correcting coding, and selection of frequency band for transmission. The same measurement is conveyed to an adaptivity function at the receive end for use in the communications receiver to select demodulator variables such as gain control, and equalization of amplitude and phase, versus frequency. The adaptivity function also permits interspersing of reverse-direction communications over the same frequency bands in a time-share mode between forward-direction and reverse-direction communication with the measurement signals having to be transmitted in only one direction. An alternate embodiment invention of this type is described which is additionally useful for mobile communications channels. Another variation embodiment is described for pure propagation measurements only, absent conveyance of end-user information.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates generally to a wirelesscommunication channel and more particularly to a channel measurementtechnique that is employed concurrent with communications.

[0003] The invention combines a novel way to realize a vector networkanalyzer (class 324/subclass 615, 607, 606) and a communications link,the former operating concurrent with the latter, in such a way thatinformation about the channel behavior is supplied directly andexpeditiously to an adaptivity function associated with operating thecommunication link over a fading channel.

[0004] Examples of adaptivity aided by this invention are, adjusting thedata rate, center frequency, type of modulation, bits per second, andforward error correcting coding.

[0005] Another adaptivity function is channel equalization (class708/subclass 323). Prior art for equalization uses one or more of thesemethods: preambles, training sequences, embedded pilot tones, receivedand use of data-aided equalization (375/subclass 229, 232). With ourinvention, these equalization preambles, training sequences and pilottones can be eliminated because the new vector network analyzer (VNA)produces the information needed to derive the correct equalizer, anddoes it better and faster than preambles, etc., with much less overhead.

[0006] Wireless unlicensed band communication and higher-poweredlicensed band communication are examples of fading channels (class455/subclass 506). What is remarkable about our invention is that thefading channel being tested by the VNA is the entire propagation path,with the ends separated by hundreds of meters to tens of kilometers. Ourinvention is a breakthrough in the state of the art because it frees upthe VNA from being merely a single piece of test equipment. Based ondiscoveries made concerning the elimination of artifacts, our inventionallows a VNA to be distributed over very long propagation paths typicalof wireless communications.

[0007] The resulting tight synergistic coupling of channel measurementwith channel usage is not possible by any other means and can lead tothe design of more efficient high speed channels for broadbandcommunications.

[0008] Because the invention involves time interlacing of shortmeasurement waveforms along with communications signals, this inventionalso has some resemblance to channel sounders (455/67.4) used both inpure research and in certain types of communications like meteor burstand HF communications. Taking meteor burst as a prior art example,channel sounding is fairly crude in the sense that it is a GO/NO-GO typeof test, either there is a meteor trail or there isn't. Our inventionsends precisely calibrated test signals time interlaced in anunobtrusive manner with communications. Some of the prior art channelsounders can measure amplitude versus frequency but cannot measure phaseshift versus frequency as this invention can. Phase shift versusfrequency is crucial to understanding the multipath behavior of thechannel which, in turn, leads to a multiplicity of adaptivity optionsnot available by other means.

[0009] The following Table 1 summarizes the field of the invention byreference to class and subclass categories as recently defined by theU.S. Patent Office. TABLE 1 Representative Prior Art: Categories Closeto This Invention Class Class Title Subclass Subclass Title 375 Pulse orDigital 229 Equalizers Communi- 231 Calibration of automatic equalizerscations 224 Testing 232 Equalizers with adjustable taps 240.02 Adaptivecoding depending on signal 324 Electrical 615 Transfer Function TypeMeasuring Characteristic Devices 607 Including A/D Converters 606Including Signal Comparison Circuit 455 Telecom- 500 Plural transmittersor receivers munications 504 Fading Compensation 506 Rayleigh orMultipath Fading Diversity Combining 505 Due to Weather 515 ControlChannel Monitoring 65 Anti-Multipath 67.4 Using a Test Signal 67.6 PhaseMeasuring 708 Electrical 300 Filters Computers: 323 EqualizersArithmetic Processing and Calculating

[0010] 2. Description of Related Art

[0011] 2.1. Previous Efforts at Overcoming Fading

[0012] Broadband wireless access networks under 5 GHz are being plannedin the U.S. to operate in licensed bands that permit high powertransmission, for example the MMDS, the ITFS, or the WCS bands, in theU.S. These bands are characterized by diffraction-mode propagation ¹ ²when the link is non-line of sight (NLOS). Diffraction is the phenomenonof waves bending around buildings and other obstacles such as foliageand rolling terrain. Attenuation also increases in a highlyunpredictable manner in NLOS. In point-to-multipoint wireless networksthere may be thousands of links which do not have a direct line-of-sightpath between them and a central hub, each seeing a slightly differentpropagation path. Finally, whenever a subscriber has been off the airfor some time it is necessary to recalibrate the link to account forminor changes such as weather conditions.

[0013] Under 5 GHz, where diffraction modes are significant ³ ⁴ ⁵ ⁶ ⁷,use of the band in a purely line of sight (LOS) mode would be wastefulof the spectrum and would require a large amount of towerinfrastructure. On the other hand, worst-case designs into the outerreaches of NLOS operation based on long-term statistical models oflinks, is also wasteful. Statistical models do not lead to results thatscale into the network capacities needed for commercial viability ofwireless Internet systems. The use of low power Industrial Scientificand Medicine (ISM) bands for broadband wireless access (BWA), whilemaking the bands readily available without cost, would also requireheavy infrastructure buildouts ⁸.

[0014] Since most broadband applications today are two-way, themeasurement should apply to both directions, adding to the complexity ofimproving performance over this type of link. The following is a summaryof various prior art adaptivity techniques which have been contemplatedfor overcoming the effect of fading in wireless channels.

[0015] (a) Adaptation Through Use of Handshake Signals

[0016] Traditionally, wire line modems adapt to the physical medium bymeans of a start-up protocol which consists of trial messages, answeringhandshakes, and modulation fallback ⁹ ¹⁰. The two modems agree on astable data rate and then begin information transfer. Wireless channelsin diffraction mode change too fast for this method to ever reachequilibrium so another method is needed which works quickly andunambiguously to define the channel's capability.

[0017] (b) Handbook-Based Designs for Fading Channels

[0018] As a design approach, handbook-based channel models are aholdover from microwave radio relay fixed point-to-point links ¹¹. Theseare typically oversized by 20 dB to 30 dB to yield very highavailability over the course of one year. The use of statistical channelmodels also finds use for mobile wireless telephony where doppler ratesto a moving vehicle are high and predictability of the fine-grainedchannel structure in real time is nearly impossible ¹².

[0019] A number of models have been developed to enable engineeringdesign to take place on a simulation basis and on a worst-case basis.Examples of these models are, Okamura, Longley-Rice, Egli, and Carey ¹³,the Ricean K-Factor model ¹⁴, and the Rayleigh fading model ¹⁵. However,reliance on handbook-based channel models for guessing what broadbandwireless channel characteristics are going to be, especially when thosemodels are more appropriate to narrow band mobile environments ¹⁶, isnot a good idea.

[0020] (c) Embedded Pilot Tones Used to Adapt to Fading Channels

[0021] Yet another approach for training a receiver to deal withpropagation conditions is the use of embedded pilot tones, specificallyrecommended in the literature for orthogonal frequency divisionmultiplexing (OFDM) ¹⁷, a modulation technique being considered bystandards groups specifically for use with broadband wirelessnon-line-of-sight links. The literature has many examples of OFDMtime-frequency occupancy plans in which certain slots are reserved forunmodulated pilot tones that step sequentially across the band ¹⁸ ¹⁹ ²⁰.

[0022] The embedded pilot tone approach suffers from the followingproblem: its determination of the channel is necessarily after-the-fact.What happens if the channel quality turns out to be too poor to supportthe data rate contained in that particular packet? Logically, it wouldbe better to know what the channel's capability is before selecting themodulation, FEC, data rate, center frequency, and so on. Pilot tonesalso take up a significant percentage of the useful information-bearingcapacity in the packet.

[0023] The invention described here overcomes another serious problemassociated with embedded pilot tones, a problem that is not well treatedin prior art systems. Since pilot tones are located at specificfrequencies, the resultant channel estimate is at best a sampled versionof the frequency response of the channel. If these samples are toowidely spaced, the frequency transfer function needs to be interpolated.Sometimes this interpolation works, other times it may not work, forexample if the delay spread in the channel is large, there could be alot of selective fading. To make the problem worse, large delay spreadis associated with high powered licensed bands where distances go up tomany kilometers.

[0024] (d) Preambles Used as Channel Sounders for Adaptivity

[0025] Yet another adaptivity mechanism for use on fading channels, inparticular to assist in finding an equalizer, is the use of a knowntraining sequence as a preamble (or in some cases a “midamble”²¹) toeach packet. This approach is typically used in Single Channel (SC)operation where selective fading causes intersymbol interference (ISI)²² ²³ ²⁴. Just as with embedded pilot tones, the channel adjustmentmechanism called into play takes place after modulation and data ratehave been selected, and so is of questionable value as a way to adapt tofading channel conditions.

[0026] (e) Diversity and Multiple Antenna Assemblies

[0027] The use of diversity techniques dates back many decades toline-of-sight radio relay systems, to HF communications, and totroposcatter. More recently diversity, and multiple input—multipleoutput (MIMO) antenna assemblies in particular, have been suggested foruse in broadband wireless systems. The inventors believe that there is alot of promise to this approach. Familiar with the extensive literatureon the topic, the inventors set out to understand what needs to be doneto implement diversity. We found out that having precise knowledge ofthe channel on each diversity path is just the first step in designing adiversity system, but by no means a guarantee of success.

[0028] A simple example is switch diversity versus equal gain combiningdiversity. With switch diversity, the strongest of several channels isselected and used. With equal gain combining diversity, all channels areused simultaneously by coherently summing together signals from severalantennas.

[0029] Which is better, switch diversity or equal gain combining?Surprisingly, it turns out that even if knowledge of the diversitychannels is perfect and both combining schemes are done correctly, thanequal gain combining may or may not be better than switch diversity,depending on the signal to noise ratio of each diversity channel. Thisapparent paradox points out the criticality of first, knowing thechannel conditions, and second, using that knowledge correctly. Thesetwo topics, measurement and adaptivity, are the subject of thisinvention.

[0030] The inventors are familiar with all of these techniques, (a)through (e), and have followed discussions in various standards settingbodies concerning the need for adaptivity in broadband fading channels²⁵. We are convinced that it is necessary to decouple channelmeasurement from communication, as this invention does, in order to havethe most freedom of adaptivity.

[0031] 2.2 Related Art for Vector Network Analyzers

[0032] A laboratory measurement of a linear system's transfer functionH(jω)†, say a filter characteristic or an amplifier pass bandcharacteristic, may be accomplished with a tool variously called anetwork analyzer, ²⁶ vector voltmeter, and vector network analyzer(VNA). Before this invention, the VNA was limited in practical use tobeing a single piece of test equipment.

[0033] Pahlavan and Levesque ²⁷ reported on using a conventional VNA formeasuring the multipath delay spread characteristic in an indoorwireless environment. In their description, a conventional VNA wasattached to the transmitter and receiver via long cables (tens ofmeters), and the VNA sweep probe was converted to a radio frequencysignal and radiated into the indoor environment while the receiver'ssignal was brought back to the output port of the VNA afterdownconversion. The cited authors were then able to compute multipathcharacteristics in an indoor environment by appropriate signalprocessing of the measured H(jω) produced by their VNA.

[0034] Note that the use of a hardware connection to a singlecentralized device is out of the question over an active wireless link,but may be suitable for modeling and statistics gathering in acontrolled indoor environment.

[0035] Getting around the problem of using cables to connect to aconventional VNA is not easy, but has been done before. In a paper byBaum²⁸ a channel modeling experiment was described in which two rubidiumfrequency standards were used, one at each end of a wireless link in theSan Francisco Bay area. They were able to find phase shift versusfrequency for this link, thanks to the stability of the rubidiumstandards.

[0036] In another study done in 2000 at Virginia Tech by Hao Xu ²⁹ aconventional VNA connected by cables to the input and output of the linkwas suggested for doing short-range line-of-sight measurements of a 38GHz wireless link multipath channel. In the same report, Xu goes on todescribe a technique called a sliding correlation system (SCS) which herecommends for long distance propagation paths. As with the Baumexperiment, Xu also uses rubidium frequency standards at each end of hisSCS link. Briefly, his idea is to let the sliding correlator findcorrelation peaks due to multipath. In turn, the multipath peaks lead toa characterization, if desired, of amplitude versus frequency and phaseshift versus frequency.

[0037] Using these two papers as a baseline, the inventors also searchedU.S. patents which mention the words network analyzer ³⁰. It appears tous that that use of rubidium standards is the best and only prior arttechnique for realizing a distributed VNA. However, what is needed toachieve commercial viability and scaling to networks with thousands ofusers is a way to accomplish the equivalent measurement usinginexpensive crystal oscillators, as well as a way to use the VNAconcurrently with communications.

[0038] The inventors would now like to make clear their reasons foremphasizing the centrality of the transfer function H(jω) in wirelesscommunications by citing some references and known facts.

[0039] The literature abundantly shows that optimal receiver processingfor any communications link, not just fading links, depends on accurateknowledge of the channel transfer function ³¹. For example, code wordEuclidean distances ³² with forward error correction (FEC) codesdetermine the ultimate bit-error-rate performance. In turn, the code'sdistance properties can be calculated once H(jω) is known. The amplitudepart of H(jω) is particularly important in establishing a link budgetand maximum possible bits per second, the phase shift part of H(jω) isparticularly important for equalization and the successful use of codedphase modulation ³³

[0040] H(jω) exists as a physical reality and can be measured. Thepreviously cited references by Pahlavan, Levesque, Baum, and Xu confirmthe fact that H(jω) is central to understanding the ability of awireless link to carry communications.

[0041] Considering the difficulty evident in the literature in goingfrom a single piece of test equipment in a laboratory to a distributedmeasurement of a propagation channel, it is worthwhile to examinecarefully what makes these situations so profoundly different. The listwhich follows takes a step-by-step approach to demonstrating theartifacts which are introduced when it becomes necessary to have twowidely separated end points, plus the use of inexpensive crystaloscillators at those end points. These are the difficulties theinventors had to overcome:

[0042] Separate local oscillator sources at transmit and receive endswill always inject some unknown phase noise process, referred to in thisApplication by φ(t), which directly cause errors in the phase angle partof the H(jω) measurement. In a single piece of test equipment one hasthe luxury of using the same oscillator at both ends of the device, sothis problem doesn't exist. Note, that φ(t) is the end-to-end phasenoise taking into account up-conversion and down-conversion to thefrequency band being used.

[0043] Separate transmit and receive systems always have some initialuncertainty regarding the local oscillator long term stability settings,characterized as Δf/f which can throw off measurement of phase. Onceagain, the laboratory network analyzer using a single frequency sourcedoes not have this problem. The Baum and Xu experiments compensated fornon-zero Δf/f by using rubidium frequency standards, carefully aligningthe equipment to effectively yield Δf/f≈0 before and during each test.

[0044] Separate transmit and receive systems always have some initialuncertainty in synchronization. This Application refers to sync error asτ_(ε). The conventional VNA's probe signal in contrast to this inventionis always in sync between input and output of the device under test.

[0045] Separate transmit and receive systems always have some initialuncertainty in the local oscillator up-conversion and down-conversionfrequencies which causes an initial uncertainty in the received signal'scenter frequency, referred to here by f_(ε).

[0046] All of these errors cause time-varying rotations in the receivedin-phase and quadrature components' complex number representation,masking the correct phase of H(jω) by large amounts in some cases,unless corrective action is taken. In this Application, the unwantedrotations are called artifacts, removal of which is described inDETAILED DESCRIPTION section 3.

[0047] The received measurement probe signal in a wireless transmissionchannel will be noisy, and may need to be filtered or processed beforeit can be useful.

[0048] Slow, stepped-frequency signals used in network analyzers aretotally inappropriate for transmission over a communications networkwith thousands of subscribers, since this signal will interfere withdata communications packets.

[0049] Subscriber equipment must be inexpensive, so theequipment-related errors described above, namely φ(t), τ_(ε), Δf/f, andf_(ε), may be very significant especially soon after the subscriberfirst turns on the receiver. The challenge faced by the inventors washow eliminate these imperfections.

[0050] 3. Objects and Advantages of this Invention

[0051] It was clear to the inventors that repetitive concurrent readingsof the amplitude versus frequency and phase shift versus frequency of afading channel are required if truly optimum communications are to beachieved over wireless channels operating in non-line-of-sightconditions. Furthermore, such measurements have to be made quickly andaccurately, up to distances of several tens of kilometers, with plentyof time left over on the channel for data, otherwise the object of theinvention would be defeated. The invention described here accomplishesthese objects in a novel way by overcoming the problems alluded toabove.

[0052] First, test signals are used that are interlaced with andseparated from data communications leaving most of the time andfrequency on the channel available for data.

[0053] Second, since the end points in the scenario setting for thisinvention could be many kilometers apart, a totally new method wasinvented to insure that the observed amplitude and phase shift versusfrequency was due only to the propagation channel, and not artifacts.

[0054] Third, the invention insures that propagation measurement isfast, accurate, and easy to broadcast back to the transmitter so thatresults are useful right away.

[0055] Fourth, the invention permits cutting down on unneeded overheadthat would ordinarily be used up by prior art devices in the form ofhandshake signals, preambles, among others. Hence the recommendedapplique form of this invention is an especially useful and synergisticcombination of desirable outcomes, none of which are possible in priorart systems.

[0056] Since none of the prior and related art squarely address theunique set of challenging problems facing broadband wireless innon-line-of-sight mode of operation under 5 GHz, specific remedies areneeded. Table 1 summarizes remedies as they will be addressed in thisApplication. TABLE 1 Comparison of Conventional Approaches and RemediesMade by This Invention Why Prior Art Adaptivity and Channel SuggestedRemedies Prior and Related Estimation Techniques Are Not Incorporated inThis Art Adequate Invention Handshake Signals Very inefficient forpoint-to-multipoint. Use a single exchange, not Overloads signalingchannel. repetitive handshake. Not scalable. One injected signalsuffices for all users. Make it easy to convey measurements back totransmit end. Modulation Fallback Not scalable Use an approach that isOverloads signaling channels. independent of modulation. Measure channelbefore Pre-selection of modulation sets may be selection of modulation,not wrong. after. May be unstable if channel changes too fast. EmbeddedTones After-the-fact of selecting modulation, not Use a measurementsignal before. outside the communication No guarantee that packet willbe received packet. correctly, may have to repeat packets. Transmitpower of the Takes away needed space data in the measurement signal willbe packet. chosen independent of the Phase coherence is lost from oneend of transmit power of the band to the other. communications. May beunstable. Preambles After-the-fact of selecting modulation, notEliminate need for preambles before. Large overhead on data. Usepropagation measurement to set equalizer directly. Transmit power oftest signal will be chosen independent of communications. ConventionalVector Two ends of test must be close together. Make endpointsindependent. Network Analyzer Always need cables, OK only for modelingand scientific studies. Vector Network Two ends of test must use veryaccurate Use crystal oscillators at the Analyzer over Long atomicfrequency standard which is often end points. Range Fading Link resetand compared. OK as model builder for simulations and Remove artifactscaused by for long-term analyses frequency errors. Cannot be usedconcurrent with Restrict measurement to only communications in low-costnetwork. bands of interest.

SUMMARY OF THE INVENTION

[0057] 1. Overview

[0058] The invention is pictured in FIG. 1. It is anapplique-combination of a novel vector network analyzer and an adaptivewireless communication link. The vector network analyzer realized inthis invention provides crucial enabling information for adaptivityfunctions at both ends of a link, raising the over-all efficiency,equilibrium and quality of service in the network in which it used.

[0059] A test signal generator 1 interlaces test signals withcommunications signals from a data modulator 4 in a burst-mode operationby means of a time-multiplexing device 15. At point 15, the power levelsare set independently. A facsimile test signal generator at the receiver2 is adjusted through the intervention of a logical processor 3 whichcould be a personal computer or could be a task-specific processor. Adownconverter 6 uses the facsimile test signal to heterodyne thereceived test signal. In turn, this downconverted signal is filtered in7, A/D converted in 8, and sent to the logical processor.

[0060] The logical processor works in either one of two modes, anacquisition mode or a tracking mode. As a rule, acquisition mode occursright after a cold start of the receiver. Tracking mode on the otherhand occurs after acquisition was successful and enables the receiver tofollow changes in the channel. If the signal is lost for any reason, thereceiver will go back to acquisition mode.

[0061] Finally a measurements conveyance part distributes thepropagation measurements to adaptivity functions at both ends.Measurement conveyance begins at the logical processor where the resultsof the measurement 33 are transferred to an adaptivity function 9 in theform of a numerical file. The adaptivity function, like the logicalprocessor, may be realized in task-specific hardware or could, forexample be part of a personal computer.

[0062] Conveyance continues by means of a data source 18 queuing thenumerical file along with possible end-user data coming from a networkinterface 25. The ultimate destination of the numerical file with thechannel measurement is the adaptivity function 34 at the transmit end.After 18, conveyance continues with data encoding 19, data modulation32, up-conversion 14, and time multiplexing over the radio frequencychannel by 17 for transmission 35 over the same band of frequencies usedby the test signal. After reception by 26, the encoded channelmeasurement is down converted in 29, filtered in 30, A/D converted in31, demodulated in 21 and routed by a demultiplex device 27 to theadaptivity function 34.

[0063] Channel measurements, having reached the adaptivity functions 9and 34, are then interpreted in 9 and in 34 for discretionary use inmodifying control parameters in the data demodulator 10, the dataencoder 19, the data modulator 32, all at the receive end of the link,and in the data encoder 20, the data modulator 4, and the datademodulator 21, at the transmit end of the link. The ability to modifycertain control parameters of modulators and demodulators in the courseof communications is well known in the art. What is remarkable aboutthis invention is the speed with which the channel information is madeavailable, the clarity and accuracy possible using the test signalsdescribed here, and the subsequent efficiency gains in trading offcommunications air time with measurement air time.

[0064] The wireless radio link represented by antennas 26 and 35, use aportion of an allocated band of frequencies for communication in bothdirections, the toggle switches 16 and 17 working in synchronism toenable the frequency band to be used in both directions. Localoscillator components 93, 87, and 86 bring the radio frequency signal toor from an intermediate frequency signal. The process of up-conversionand down-conversion in 28, 29, 88, 13, and 14, along with poweramplification in 93 and 90, low noise amplification in 89 and 92, areall conventional, and well known techniques used by radio frequencydesigners of data communication front-ends, going by the name,time-division-duplex (TDD). What is new and significant from an economypoint of view in a large network, is that the test signal is needed inonly one direction. The invention purposely takes advantage not only ofduality in the external physical properties of wireless propagationlinks, but also in duality and stability of the internal circuitry. Inparticular the fact that the propagation is being measured between twopoints internal to the invention circuitry, point 96 and point 95, andnot between two external points, is critical to the operation of theinvention.

[0065] The description given above for conveyance of a numerical filefrom the logical processor to a distant station can be considered anormal piggy-back function of a wireless communications link. What isnew about this conveyance is first, that the numerical file with channelinformation is being conveyed before the channel fading has had a chanceto change, and second, the numerical information is available and usedto update settings in the communications portions of the wireless linkat both ends of the link in an expeditious fashion.

[0066] Hence, it will be understood that the operation of the VNAfunction does not depend on drawing a distinction between communicationcoming from the end-user's network interfaces 22, 23, 24, and 25, andcommunications internally generated by the measurements and conveyancefunctions, once the measurements are piggy-backed. When communicationsis coming from or to a network interface 22, 23, 24, 25, it will bereferred to here for clarity as end-user data, while if thecommunications is internally generated to convey information about thechannel and subsequent adjustments to be made in the data encoders, thedata modulators and the data demodulators, as noted above, then it willbe referred to as internal control information.

[0067] The preferred embodiment for the test signal is a composite ofsegments of linear FM sweeps, one of which is shown in FIG. 2. Thein-phase and quadrature functions of a linear FM sweep 37 are obtainedfrom equation (1) in DETAILED DESCRIPTION, section 2. An FM sweep signalhas a start time 38 and a stop time 39 and a slope 40 whose units are Hzper second, and a power level. FIG. 3 shows settable parameters assumedfor the generators at both ends: frequency band of interest 43, timeduration of a specific sweep 47, rate of injection 41 and 42, and theslope 46 of either an upsweep or downsweep or both. Note, that some ofthese settable parameters are interdependent: slope depends on frequencyspan of interest 43 and duration 47 of sweep. The facsimile generatorhas three additional adjustable control parameters shown in FIG. 4,frequency offset 48 from nominal, time offset 49 from nominal, and timebase 50. Note, that these additional three parameters are all withrespect to the settable parameters so an actual realization of afacsimile test signal may look very different from that shown in FIG. 4.FIG. 5 shows two examples of preferred embodiment variations in the waytest signals are interlaced with communications.

[0068] The band of interest 51 or 59 in FIG. 5 is purposely left ingeneric form in the description because the invention can be used in avariety of situations where the band of interest changes. MMDSallocations in the United States come in quanta of 6 MHz. WCS bands are5 MHz. Test signals 52 are shown in a specific realization ascombinations of two upsweeps 53 and 55 with downsweep 54. As a matter ofdesign discretion, this test signal could just as well have been twodownsweeps and one upsweep. An interlaced set of test signals is alsoshown at 58 as a repetition of 52. The fact that 52 contains threesegments is related to the solution of the equations for artifactremoval. Acceptable variations to the preferred embodiment can be foursweep segments per interlace, five sweep segments per interlace, etc.There is no prescribed number that must occur per interlace.

[0069] Example 2 of FIG. 5 shows a variation in how test signals can getinterlaced with communications: instead of just forward-directioncommunication, forward communications 62 is here time multiplexed with areversed direction communication burst 63, occupying the same frequencybands. This radio frequency multiplexing operation is carried out bysynchronized toggle switches 16 and 17. Note that for illustrationpurposes, 62 and 63 are shown striated into 12 frequency channels each,64 and 65. The number of channels shown, 12, in 64 and 65 is purely forexample purposes and does not represent a restriction on how theinvention is used. The point being made here is that the sweep bandwidthand the communications bandwidth of an individual end-user, do not haveto be the same. The decision on channelization, length of burst, use ofreverse channel communication, are all situation-dependent.

[0070] It is now useful to point out, after seeing FIGS. 3, and 5, thatthere is an interplay of design choices possible between the settableparameters of the test signal and the use of interlaced communications.It should be clear to someone familiar with designing the air interfacefor a communication network that proper use of the settability of thetest signal permits efficient interlacing, efficient in this casemeaning using up very little of the actual air time, leaving most of theair time available for communications. The inventors discovered that thetime segments devoted to test signal, 51, 58, 60, and 66, for example,can be made just a few percent of the available air time withoutsacrifice of accuracy of measurement.

[0071] As just noted, settability is used as a mechanism to flexiblyinterlace test signals and communications. Once settability parametersare chosen, they may stay fixed for long periods, say during aparticular communications transaction, and in fact may never change atall in a mature design. On the other hand, the adjustable controlparameters on the facsimile signal shown in FIG. 4 are used in afeedback control loop created in the logical processor 3 controlling thefacsimile generator 2 which performs the heterodyne operation 6, so tendto get used on a quasi-continuous basis during normal operation.

[0072] Notable in this invention is the presence of the third controlparameter 50, time base. The inventors realized that one of severalartifacts introduced into the vector analyzer measurement came about dueto frequency instability of the crystal oscillators. This artifact, dueto the relative error Δf/f in oscillators, is a phase versus frequencyadditive component that varies parabolically in time, as shown inDETAILED DESCRIPTION, section 3. The ability to vary the time base 50 ofthe receiver facsimile signal permits finding and eliminating thisartifact.

[0073] 2. Concept of Operation

[0074] The inventors envision that this VNA applique will be part of anopen standard interoperable with current and evolving air interfacestandards for broadband wireless systems. We believe that it isessential to broadband wireless evolution and wish to influencestandards-setting bodies to eventually adopt one or more of the manypossible preferred embodiment variations, as part of emerging PhysicalLayer (PHY) specifications. For this reason, modulation, forward errorcorrection coding, etc., of the communications has been purposely leftgeneric in this invention description.

[0075] Legacy PHY portions of modems are nearly 100% reusable whenchannel state information is gathered using the invention describedhere. A link which has been measured by using the vector networkanalyzer applique can be made to behave less like a fading channel andis more stable and reliable. Newer radios, as yet undeveloped, can bedesigned to more fully take advantage of the inundation of knowledgeabout channel states that will be possible with the vector networkanalyzer applique. For example, turbo coding ³⁴ performance is verysensitive to channel state knowledge. The invention therefore makesturbo coding more reliable and easier to build.

[0076] Based on this discussion, the inventors stress that there is aclean dividing line between the invention itself and the way the iteventually gets used. FIG. 6 illustrates the context-sensitivity of theinvention operations. This figure is called a ladder diagram, and iscommonly used in describing air interface processes in wireless systems.It shows in time sequence running from top to bottom of the figure, whattypical operations come first, second, and so on, and some of thetypical branching steps that are possible, depending on channelconditions and measurement outcomes. FIG. 6 is by no means an exhaustivelisting of the possible operations of the invention.

[0077] The way the vector network analyzer applique would work inpractice is that at the start of a communication session, which could bean Internet TCP/IP exchange, the test signal generator node 68 in FIG.6, referring back to 1 in FIG. 1 would begin broadcasting test signalsto all listeners who would in one possible realization be connected tothe transmit end in a hub-spoke configuration, in another possiblerealization might be just a single transmit-receive pair. The followingtable relates FIG. 6 action nodes along the top of the figure tocomponents shown in FIG. 1. TABLE 2 Relating Nodes in Ladder Diagram toInvention Components Shown in FIG. 1 Reference number Compositereference Action node in FIG. 6 used in FIG. 6 used in FIG. 1 Datasource at transmit end 67 22 Test signal generator 68 1 Adaptivityfunction at 69 34 transmit end Control message generator 70 34 & 5Facsimile generator 71 2 Logical processor 72 3 Adaptivity functions at73 9 receive end Control channel 74 33 & 9 & 18 measurement conveyanceControl message receiver 75 10 & 36 & 9  Receive end comm 76 arrowleading into 10 adjustment input coming from 9 Receiver for data 77 24

[0078] These test signals would initially be interlaced with broadcastcontrol signals only, indicating to the receiver that the transmit endwas ready to carry on a conversation. The listener's initialresponsibility would be to acquire the test signal 78, transfer fromacquisition mode to track mode 79, then go into a communications mode 85as well as a concurrent measurements conveyance mode 82. Aquasi-steady-state condition of the transmitter and receiver would bereached when both ends were continuously communicating with each other85 while carrying out 79, 82, 83, and 84 in a concurrent mode.

DRAWINGS

[0079]FIG. 1

[0080] A block diagram of the VNA applique invention.

[0081]FIG. 2

[0082]2(a) is a plot of the linear FM sweep signal's in-phase andquadrature components. In the preferred embodiment, the test signal ismade up of segments of linear FM sweeps.

[0083]2(b) is a plot of the frequency versus time in an upward-goinglinear FM sweep.

[0084]FIG. 3

[0085]3(a) is a sketch showing two settability characteristics of thetest signal: variation in frequency range and variation of repetitionrate.

[0086]3(b) shows three additional settability parameters: time spacingbetween successive up and down sweeps, slope in Hz per second of thelinear FM sweep, and time spacing between successive down and up sweeps.

[0087]FIG. 4

[0088] Three control parameters of the facsimile signal: frequencyoffset, time offset, and time base.

[0089]FIG. 5

[0090]5(a) is an alternative embodiment showing three discrete andtime-separated measurement sweeps interlaced with forward-directioncommunications.

[0091]5(b) is an alternative embodiment with both forward-direction andreverse-direction communication interlaced with measurement testsignals. A possible communications option, that of channelization isillustrated in this figure.

[0092]FIG. 6

[0093] A ladder diagram illustrates one possible context of theinvention's use, showing that it is closely integrated with start-up andcontinuing operation of a communications session on a wireless network'sair interface.

[0094]FIG. 7

[0095] Illustrates gross misalignment of the test signal and facsimileafter a cold start.

[0096]FIG. 8

[0097] A sketch of the condition just after acquisition is complete,where the difference frequency between the transmitted and facsimilesignal lies within the filter 7 bandwidth, see also FIG. 1.

[0098]FIG. 9

[0099] A block diagram of the heterodyne operation at the receiverbetween the received signal and the facsimile signal, when there ismisalignment of the test signal and facsimile signal soon after coldstart. This condition exists for a short period of time before theacquisition process begins.

[0100]FIG. 10

[0101]10(a) shows crossovers between the test signal and the facsimilesignal in free-running condition soon after cold start.

[0102]10(b) shows the occurrence of a coincidence, defined as aslowed-down version of a crossover that occurs when the facsimile signaltime base is speeded up. Note, slowing down the time base would alsoproduce a similar coincidence.

[0103]FIG. 11

[0104] A block diagram of the heterodyne operation at the receiver whenthe logical processor is in tracking mode, showing the occurrence ofsignal rotations proportional to frequency difference between the testsignal and the facsimile signal during an upsweep segment or during adownsweep segment.

[0105]FIG. 12

[0106] Shows how the slope of the linear FM sweep signal increases forpositive values of Δf/f due to a combination of two effects: shortertime interval for a sweep segment, and overshoot in maximum frequencydeviation of the linear FM sweep.

[0107]FIG. 13

[0108] Used in Table 6 to show the introduction of artifacts due tosmall errors in time offset, frequency offset, and oscillator offset.Artifact removal is based on taking sums, differences, and seconddifferences of the frequency separations measured by the logicalprocessor when it observes a typical progression illustrated in 13(c).

[0109]FIG. 14

[0110] Acceptable alternative embodiments of the test signal which canbe made mathematically equivalent to the triangular sawtooth FM sweepsignal of FIG. 13. Variations sketched in this figure permit room forinterlacing of communications without changing the fundamentalartifact-removal feature of this invention.

[0111]FIG. 15

[0112] Prototype vector network analyzer built and tested in 2001. Thenext five figures show plots of numerical files contained in the logicalprocessor at various stages of acquisition and track. In the prototype,the logical processor was a PC.

[0113]FIG. 16

[0114] Appearance of the coincidence signal in the prototype caused byspeeded up facsimile signal heterodyning a received signal duringacquisition mode.

[0115]FIG. 17

[0116] Same as FIG. 16 with expanded time base. Shows in-phase andquadrature signals during a coincidence as seen in the logicalprocessor.

[0117]FIG. 18

[0118] Signal seen by logical processor in tracking mode during a singledownsweep of the linear FM signal. Artifact due to Δf/f is stillpresent, resulting in a parabolic phase versus time which is visible inthis figure.

[0119]FIG. 19

[0120] After removal of the parabolic phase versus time artifact andsetting frequency offset to zero, the logical processor outputs thissignal combination, which has both an in-phase and quadrature signalcomponent, as a time analog of H(jω) over the sweep bandwidth which inthe case of the prototype constructed in 2001, is approximately 6 MHz.

[0121]FIG. 20

[0122] Same numerical file as in FIG. 19 but plotted with in-phase onx-axis against quadrature on y-axis, to create a polar plot of H(jω).

[0123]FIG. 21

[0124] Example of an alternative embodiment with CDMA instead of linearFM sweep test signals.

[0125]FIG. 22

[0126] Controllable parameters on an alternative embodiment CDMAfacsimile signal are, time offset, frequency offset, and time baseoffset.

[0127]FIG. 23

[0128] Alternative embodiment of the vector network analyzer as a puremeasurement system instead of a communications applique system. ThisFigure repeats the numbering system used in FIG. 1 and eliminatescertain end-user interfaces which are no longer needed.

DETAILED DESCRIPTION

[0129] 1. Preferred Embodiment

[0130] In this description, the preferred embodiment uses test signalsthat are segments of linear FM sweep signals. Many combinations of suchsignals are possible. The nature and benefits of the invention are notdependent on the precise format and placement of linear FM sweepsignals, so there may be many variations in how the sweep signals areinterlaced with communications. Additional embodiments such as the useof CDMA test signals are also possible and will be described in section5.

[0131] The underlying purpose and benefit of the invention is itsability to compute, distribute and make use of, a frequency transferfunction for each individual link in a wireless network subject tofading channel conditions.

[0132] At the start of the procedure envisioned for any typical use ofthe invention, there is a two-step process called acquisition andtracking which is independently performed in each receiver while atransmitting station broadcasts test signals interlaced withcommunication.

[0133] Acquisition

[0134] A transmitted test signal is sent out by one station and receivedindependently by a number of receiving stations. Since the receivingstations begin operation from a cold start, their knowledge of thetransmit signal's timing is nil to begin with, so a boot-up type ofoperation is needed whereby each receiver sequentially discovers variousfacts about the transmitted signal that allows it to reduce themisalignment. The presence of gross misalignment at start-up isillustrated in the sketch in FIG. 7.

[0135] Completion of the acquisition step, however close it may be, isstill not perfect alignment. The next mode, tracking, is meant to reduceany remaining misalignment errors to zero.

[0136] Tracking

[0137] As just noted, the tracking stage commences when acquisition hasreduced misalignment errors to a sufficiently small condition.Conditions during tracking are illustrated in the sketch in FIG. 8, andcan be characterized as the maintenance of a small difference frequency,small enough to keep the downconverted signal within a relatively narrowbandwidth of the filter preceding the A/D converter in FIG. 1, filter 7.

[0138] Tracking is a continuous process that exists as long as thereceived signal strength is sufficient. If the signal is lost,acquisition may again be necessary. In the preferred embodiment,adjustments are made during tracking mode by the logical processor,abruptly in the form of a correction which eliminates all the artifactsat once. The logical processor then has available the desired channelestimate in numerical form. A prototype version discussed in section 4performed the artifact removal as a continuous feedback operation forreasons which are given in that section. Continuous feedback takeslonger than the preferred embodiment.

[0139] In summary, the new VNA works in a two-step process, acquisitionand tracking, during which equipment imperfections such as timing error,frequency offset error, among others, are progressively reduced andeliminated. The logical steps used in this invention are listed in Table3. TABLE 3 Logical Steps During Acquisition and Tracking Modes Stage ofOperation step # Receiver Processing Notes and Explanation Cold Start 1Turn on receiver. It may be necessary to return to this step in theevent the signal is lost, or at Begin free-running generation of thestart of a new communications facsimile signal. transaction. Acquisition2 A/D convert filtered baseband signal FIG. 9 illustrates the heterodyneobtained by heterodyning received signal operation with free runningfacsimile. with locally generated replica. Crossover points create shortpulses which are not easily detectable. 3 Switch time base of facsimileto FAST or Uses technique shown in FIG. 10(b). to SLOW. Logicalprocessor searches Logical processor controls time base to forcoincidence. Logical processor create two coincidences. Location ofcommands facsimile to alter start time coincidences determines removalof and time base. Search for second gross misalignment errors.coincidence. 4 Correct timing and frequency offset in When this step issuccessful, the locally generated replica of transmit acquisition modeis over and signal. This is an abrupt correction that heterodyned signalis confined to the should bring the downconverted signal bandwidth ofthe filter 7 preceding the within the baseband filter. Facsimile A/Dconverter 8 of FIG. 1, as time base reverts to nominal. sketched in FIG.8. Tracking 5 Examine output of baseband filter. Was Frequencydifference between the acquisition successful? NO: go back to receivedand locally generated signal step #3. YES: continue with step #6. nowtotally within filter 7 bandwidth. 6 Count rotations on successiveupsweeps Counting rotations is the preferred and downsweeps. embodiment:rotation counting is a specific realization of a frequency estimate.FIG. 11 sketches heterodyne operation and rotation counting tracking. 7Use counts to correct artifacts. Intermediate stage in tracking.Alternative embodiment of this step is to Preferred embodiment goesdirectly to use feedback control loop to gradually step #8. eliminateartifacts. 8 Force timing error to zero. Explanation is given in section3 for Force frequency offset error to zero. artifact removal. Uses sumand Force oscillator instability to zero. differences of rotation countsto solve for and to remove artifacts, TABLE 6. 9 Calculate H(jω). Thisfunction, which is the desired output of the new VNA, is contained in afile in the logical processor after artifact removal, as shown inequation (19).

[0140] Steps 2, 3, and 4 in Table 3 will now be described in somewhatmore detail. Since the free-running probe generator in the receiver maybe relatively far off in timing and in frequency offset, the incomingsignal and the locally generated signal will repeatedly cross, producingshort periodic pulses into the baseband filter, as shown in the sketchin FIGS. 9 and 10(a). These pulses will not be sufficient to produce aclear determination of the timing and frequency offsets. The conceptadapted by this invention to facilitate acquisition is to make thelocally generated signal artificially fast or artificially slow so thata relatively long, higher powered, coincidence pulse occurs when the twosignals overlap, as sketched in FIG. 10(b).

[0141] It should be clear that it does not matter whether the signalruns fast or slow: it is always going to be easier to detect theartificial coincidence than to detect periodic crossovers.

[0142] Having measured the coincidence on (in this case) the downsweep,the locally generated signal can be adjusted so that a coincidence nextoccurs on the upsweep. With two such measurements, the timing andfrequency offset error can be computed and the locally generated signalcan be adjusted again so that the difference frequency lies entirelywithin the filter 7 bandwidth. Then step #6 in Table 3, the trackingstage of the measurement, begins.

[0143] 2. Calculation of H(jω)—Ideal Case No Artifacts

[0144] Probe signal building blocks used in the preferred embodiment aremade up of what is variously called a chirp signal, a frequency sweepsignal, or linear FM signal, see FIG. 2. The complex (bandpassequivalent) of a chirp signal or linear frequency modulated signal is ³⁵³⁶ ³⁷: $\begin{matrix}{{s(t)} = {{^{{j2}\quad \pi \frac{1}{2}{mt}^{2}}\quad {for}{\quad \quad}0} \leq t \leq T_{FM}}} & (1)\end{matrix}$

[0145] Note, that when the phase angle of the complex signal isdifferentiated, one gets the instantaneous angular frequency as afunction of time: $\begin{matrix}{\overset{.}{\theta} = {{\frac{\quad}{t}\left( {2\quad \pi \frac{1}{2}{mt}^{2}} \right)} = {2\quad \pi \quad {mt}\quad {rads}\quad {per}\quad \sec}}} & (2)\end{matrix}$

[0146] This function is a frequency ramp with slope m Hz per second.$\begin{matrix}{{f(t)} = {\frac{\overset{.}{\theta}}{2\quad \pi} = {{{mt}\quad {Hz}\quad 0} \leq t \leq T_{FM}}}} & (3)\end{matrix}$

[0147] The maximum frequency at the end of the ramp is thereforemT_(FM). FIG. 2(a) shows a time plot of the real and imaginary part ofequation (1) which become the in-phase and quadrature components of thereal signal, along with a plot of the instantaneous frequency, shown in2(b).

[0148] The equivalent impulse response channel from tower to receiver isgenerally represented as a complex bandpass impulse response, h(t) whichlasts from t=0 to some initially unknown t=T_(max), called the delayspread of the channel. A line of sight (LOS) link ideally has nodispersion.

h(t)=δ(t) in LOS: a unit impulse, delay spread is zero.  (4)

[0149] The response of the channel to the chirp signal is, byconvolution, $\begin{matrix}{{r(t)} = {\int_{- \infty}^{+ \infty}{{s(\tau)}{h\left( {t - \tau} \right)}\quad {\tau}}}} & (5)\end{matrix}$

[0150] Substituting (1) into (5), and considering the output of thechannel only inside the chirp interval T_(max)≦t≦T_(FM), $\begin{matrix}{{r(t)} = {{\int_{0}^{T_{FM}}{^{{j2}\quad \pi \frac{1}{2}m\quad \tau^{2}}\quad {h\left( {t - \tau} \right)}{\tau}\quad {for}\quad T_{\max}}} \leq t \leq T_{FM}}} & (6)\end{matrix}$

[0151] With a change of variables this integral becomes, $\begin{matrix}\begin{matrix}{{{{letting}\quad x} = {\tau - t}},} \\{{r(t)} = {\int_{- t}^{T_{{FM}^{\quad} - t}}{^{{j2}\quad \pi \frac{1}{2}{m{({x + t})}}^{2}}\quad {h\left( {- x} \right)}{x}}}} \\{= {^{{j2\pi}\frac{1}{2}{mt}^{2}}{\int_{- t}^{T_{{FM}^{- t}}}{^{{j2}\quad \pi \frac{1}{2}{mx}^{2}}\quad ^{{j2}\quad {\pi mxt}}{h\left( {- x} \right)}{x}}}}}\end{matrix} & (7)\end{matrix}$

[0152] Now recognize that the upper limit can be replaced by zero.Rearranging terms, $\begin{matrix}\begin{matrix}{{{r(t)}^{{- {j2}}\quad {\pi m}\frac{1}{2}t^{2}}} = {\int_{- t}^{0}{^{{j2}\quad {\pi m}\frac{1}{2}x^{2}}\quad ^{{j2}\quad {\pi {mxt}}}{h\left( {- x} \right)}{x}}}} \\{{{{Letting}\quad \omega^{\prime \quad}}\overset{\Delta}{=}{2\quad {\pi {mt}}}},} \\{{{r(t)}^{{- {j2}}\quad {\pi m}\frac{1}{2}t^{2}}} = {\int_{- t}^{0}{{h^{\prime}\left( \quad {- x} \right)}^{j\omega\prime x}{x}}}} \\{= {{\int_{0}^{t}{{h^{\prime}(x)}^{- {j\omega\prime x}}\quad {x}\quad {where}\quad {h^{\prime}(x)}}}\overset{\Delta}{=}{^{{j2}\quad {\pi m}\frac{1}{2}x^{2}}{h(x)}}}}\end{matrix} & (8)\end{matrix}$

[0153] Replace the upper limit by ∞ since t is always greater than thedelay spread,${{r(t)}^{{- j}\quad {\pi m}\frac{1}{2}t^{2}}} = {{\int_{0}^{\infty}{{h^{\prime}(x)}^{{- j}\quad {\omega'}x}\quad {x}}}\overset{\Delta}{=}{{H^{\prime}\left( {j\quad \omega^{\prime}} \right)}.}}$

[0154] The left hand side of equation (8) is a processed (deswept)version of the received signal. The right hand side of equation (8) is atime-analog of a close approximation to H(jω) as demonstrated next.

[0155] The delay spread, T_(max), is expected to range from nanosecondsfor LOS to 10's of microseconds for NLOS users 20 to 40 kilometers awayfrom the tower. Duration of the sweep should therefore be made muchlonger than the expected size of the delay spread. Note from (8) that asm gets smaller, the approximation for H(jω) gets better, as would beexpected. Nevertheless, in this application, we do not have the luxuryof letting m→0, so additional signal processing may be needed. Table 3shows how the impulse response becomes distorted for non-zero slope.

[0156] If the sweep bandwidth is large enough to resolve individualreflection coefficients in the impulse response, then correcting thiserror is relatively straightfoward since the correction can be doneindependently at each coefficient. Table 4 indicates that for sweeps onthe order of a tenth of a second and bandwidths of 2 MHz, this error isnot significant at all, so will be neglected in what follows. TABLE 4Distortion Due to Non-Zero Slope (Fast Sweep) In Channel Probe sweepduration delay spread 20,000 30,000 40,000 50,000 60,000 70,000 80,00090,000 100,000 μ-sec sweep 2 MHz  5 0.5 0.3 0.2 0.2 0.2 0.1 0.1 0.1 0.110 1.8 1.2 0.9 0.7 0.6 0.5 0.5 0.4 0.4 15 4.1 2.7 2.0 1.6 1.4 1.2 1.00.9 0.8 20 7.2 4.8 3.6 2.9 2.4 2.1 1.8 1.6 1.4 25 11.3 7.5 5.6 4.5 3.83.2 2.8 2.5 2.3 μ-sec sweep 6 MHz  5 1.4 0.9 0.7 0.5 0.5 0.4 0.3 0.3 0.310 5.4 3.6 2.7 2.2 1.8 1.5 1.4 1.2 1.1 15 12.2 8.1 6.1 4.9 4.1 3.5 3.02.7 2.4 20 21.6 14.4 10.8 8.6 7.2 6.2 5.4 4.8 4.3 25 33.8 22.5 16.9 13.511.3 9.6 8.4 7.5 6.8 μ-sec

[0157] Therefore, the deswept signal is in fact a replica of the channeltransfer function: $\begin{matrix}\begin{matrix}{{{deswept}\quad {signal}} = {{{r(t)}^{{- {j2}}\quad {\pi m}\frac{1}{2}t^{2}}} = {H\left( {j\quad \omega^{\prime}} \right)}}} \\{{{where}\quad \omega^{\prime}} = {{2\quad {\pi {mt}}{\quad \quad}{and}\quad T_{\max}\quad {and}\quad T_{\max}} \leq t \leq T_{{FM}_{\quad}}}}\end{matrix} & (9)\end{matrix}$

[0158] Equation (9) is the preferred embodiment logical processorcomputation in the ideal case where there are no artifacts.

[0159] 3. Non-Ideal Case Removal of Artifacts

[0160] A closer look will show how initial misalignment due to equipmentand measurement uncertainties adds signal processing burden to the taskover-and-above equation (9). The derivation leading to equation (9)assumed the end-state (step 9 of Table 3) of acquisition and tracking,namely a perfectly in-sync receiver with perfect frequency translationand perfect oscillator match at transmit and receive stations, and nophase noise. Therefore let's back up from this condition by introducingthe errors that must be discovered and removed during the trackingphase. Introducing misalignment error terms into a modified version ofequation (8) shows what happens to the measurement while tracking stepsare in operation as given in Table 3, steps 6, 7, and 8.

[0161] Let φ(t) be the phase noise in the receiver referenced to thetransmitter, from end-to-end.

[0162] Let Δf/f be the oscillator long-term instability at the receiverrelative to the transmitter.

[0163] Let f_(ε) be the initial up- down-conversion offset error.

[0164] Let τ_(ε) be the initial time sync error.

[0165] The probe signal is still the same, see equation (1), but thedesweeping signal is modified by these errors. $\begin{matrix}{{{desweep}\quad {signal}} = {^{{{- {j2}}\quad \pi \frac{1}{2}{({m + {\Delta \quad m}})}{({t - \tau_{ɛ}})}^{2}} + {{j2}\quad \pi \quad {f_{ɛ}{({t - \tau_{ɛ}})}}} + {j\quad {\varphi {(t)}}}}}_{\quad}} & (10)\end{matrix}$

[0166] Slope error Δm is caused by the relative long-term frequencyuncertainty of the receiver oscillator relative to the transmitteroscillator, Δf/f. Say Δf is positive. This not only speeds up the sweep,it causes the maximum sweep frequency to overshoot. These two effectsmultiply to give, $\begin{matrix}{\frac{m + {\Delta \quad m}}{m} = {\left( {1 + \frac{\Delta \quad f}{f}} \right)^{2} \cong {1 + {2\Delta \quad \frac{f}{f}}}}} & (11)\end{matrix}$

[0167]FIG. 12 is a sketch explaining equation (11). Therefore, thenon-ideal desweep signal in terms of initially unknown equipmentuncertainties becomes, $\begin{matrix}{{desweep} = ^{{{- {j2}}\quad \pi \frac{1}{2}{({m + {2\frac{\Delta \quad f}{f}m}})}{({t - \tau_{ɛ}})}^{2}} + {{j2\pi}\quad {f_{ɛ}{({t - \tau_{ɛ}})}}} + {{j\varphi}{(t)}}}} & (12)\end{matrix}$

[0168] Expand the square term:

(t−τ _(ε)) ² =t ²−2tτ _(ε)+τ_(ε) ²  (13)

[0169] Now collect terms and ignore second-order errors: $\begin{matrix}{{desweep} = ^{{{- {j2\pi}}\frac{1}{2}{mt}^{2}} - {{j2\pi}\frac{\Delta \quad f}{f}{mt}^{2}} + {{j2\pi m\tau}_{ɛ}t} + {{j2\pi}\quad f_{ɛ}t} + {{j2\pi}\quad f_{ɛ}\tau_{ɛ}} + {{j\varphi}{(t)}} - {{j2\pi}\frac{1}{2}{m\tau}_{ɛ}^{2}}}} & (14)\end{matrix}$

[0170] Combining all of the error terms, the processed time function atthe receiver, instead of being the wanted analog of the channel transferfunction becomes: $\begin{matrix}{{{deswept}\quad {r(t)}} = {{H\left( \omega^{\prime} \right)}^{{{- {j2\pi}}\frac{\Delta \quad f}{f}{mt}^{2}} + {{j2\pi}\quad f_{ɛ}t} + {{j2\pi}\quad \tau_{ɛ}{mt}} + {{j\varphi}{(t)}} + {{j \cdot {constant}}\quad {phase}\quad {term}}}}} & (15)\end{matrix}$

[0171] where H(ω′) was previously defined in equation (8). Collectingterms inside the exponent: $\begin{matrix}{{{deswept}\quad {r(t)}} = {{H\left( \omega^{\prime} \right)}e^{{{- j}\quad 2\pi \frac{\Delta \quad f}{f}{mt}^{2}} + {{{j2\pi}{({f_{ɛ} + {\tau_{ɛ}m}})}}t} + {{j\varphi}{(t)}} + {{j \cdot {constant}}\quad {phase}\quad {term}}}}} & (16)\end{matrix}$

[0172] Thus, f_(ε)+τ_(ε)m is a combined frequency-time error noticeableas a rapid spin in deswept r(t), rapid compared to the other effectsinside the exponent. A rough estimate of the number of cycles of spinduring a sweep time T_(FM) of the received signal is,

cycles of spin in deswept r(t)≈(f_(ε)+τ_(ε)m)T_(FM)  (17)

[0173] Meanwhile, the parabolic term${- {j2\pi}}\frac{\Delta \quad f}{f}m\quad t^{2}$

[0174] is adding a phase error which increases slowly at first, thenmore rapidly as the sweep ends. The spreadsheet in Table 5 gives someidea of the accumulated phase error due just to the parabolic term whenthe oscillator instability is comparable to commercial grade crystaloscillators. The numbers in the spreadsheet, divided by 360, give theadditional number of rotations caused by the parabolic term in (16).Order-of-magnitude, the parabolic term is not significant to the spinterm but could be very significant compared to the phase noise and tothe actual phase of H(jω). Now note from Table 4, that the parabolicterm just by itself would mask the true angle versus frequencycharacteristics of H(jω). TABLE 5 Accumulated Parabolic Phase Error Dueto Δf/f of Commercial Grade Crystal Oscillators sweep duration delta f/f20000 30000 40000 50000 60000 70000 80000 90000 100000 μ-sec sweep 2 MHz1.00E-07 1.4 2.2 2.9 3.6 4.3 5.0 5.8 6.5 7.2 5.00E-07 7.2 10.8 14.4 18.021.6 25.2 28.8 32.4 36.0 1.00E-06 14.4 21.6 28.8 36.0 43.2 50.4 57.664.8 72.0 5.00E-06 72.0 108.0 144.0 180.0 216.0 252.0 288.0 324.0 360.01.00E-05 144.0 216.0 288.0 360.0 432.0 504.0 576.0 648.0 720.0

[0175] A person familiar with OFDM modulation in general and coded phasemodulation in particular, will realize that these errors are notacceptable and must be removed. Let us now consider the logicalprocessor calculations to remove artifacts. Note, that the currentassumption, for simplicity is that the test signal is a sawtoothwaveform produced by repeating the upsweep and downsweep in a periodicfashion without any breaks in time. Without writing out all the terms,suffice it to say that the resulting sawtooth FM signal will have thesecharacteristics:

[0176] The number of cycles of spin on the first up-sweep isapproximately (f_(ε)+τ_(ε−up−1)m)T_(FM), where τ_(ε−up−1) is the syncerror on the first up-sweep.

[0177] The number of cycles of spin on the first down-sweep is,similarly (f_(ε)−τ_(ε−down−1)m)T_(FM).

[0178] When Δf/f is not zero, the sawtooth FM signal walks with respectto the incoming received signal as shown in FIG. 13(a), in such a waythat time sync keeps slipping by an amount (Δf/f)T_(FM) every T_(FM)seconds. Thus, since the original time error is designated τ_(ε−up−1) onthe first up-sweep, during the first down-sweep, sync error will have anadded amount,

τ_(ε−down−1)=τ_(ε−up−1)+(Δf/f)T _(FM).  (18)

[0179] This term keeps accumulating as shown in FIGS. 13(b) and 13(c).Table 6 shows what would be recorded if one measured the successivedifferences in cycle spins, from all sources, neglecting the smallnumber of spins contributed by the parabolic term and H(jω) itself. Itshows that the second differences in number of up- and down-spins isproportional to Δf/f and no longer contains the terms f_(ε or τ)_(ε−up−1). This fact means that by adding a minimum of two more FMsweeps, one down and one up, the error Δf/f can be separated out fromthe other two and can be computed. Once the spin components due to Δf/fare removed (set Δf/f equal to zero in Table 6), the alternating firstdifferences all become proportional to the initial error τ_(ε−up−1), sothis error may also be computed. Finally, if instead of firstdifferences, one recomputed first sums, that is 1+2, 2+3, 3+4, and so onwith Δf/f set to zero, these would all be proportional to f_(ε), so itcould then be computed.

[0180] What this brief discussion has shown is that three sweeps aresufficient to compute and therefore abruptly remove, timing, frequencyoffset, and oscillator stability errors. TABLE 6 First and SecondDifferences of Spin Counts Permit Removal of Artifacts Cycle SpinsDuring Sweep with Accumulating Delay Alternating First Which Sweep ?Error Difference Second Difference 1 Up number 1 (f_(ε) +τ_(ε−up−1)m)T_(FM) 2 Down number 1 (f_(ε) − (τ_(ε−up−1) +(Δf/f)T_(FM))m)T_(FM) 1 1-2 (2τ_(ε−up−1)m + (Δf/f)T_(FM)m)T_(FM) 3 Upnumber 2 (f_(ε) + (τ_(ε−up−1) + 2(Δf/f)T_(FM))m)T_(FM) 2 3-2 2-1(2τ_(ε−up−1)m + 3(Δf/f)T_(FM)m)T_(FM) 2(Δf/f)T_(FM) ² m 4 Down number 2(f_(ε) − (τ_(ε−up−1) + 3(Δf/f)T_(FM))m)T_(FM) 3 3-4 3-2 (2τ_(ε−up−1)m +5(Δf/f)T_(FM)m)T_(FM) 2(Δf/f)T_(FM) ² m 5 Up number 3 (f_(ε) +(τ_(ε−up−1) + 4(Δf/f)T_(FM))m)T_(FM) 4 5-4 4-3 (2τ_(ε−up−1)m +7(Δf/f)T_(FM)m)T_(FM) 2(Δf/f)T_(FM) ² m 6 Down number 3 (f_(ε) −(τ_(ε−up−1) + 5(Δf/f)T_(FM))m)T_(FM) 5 5-6 5-4 (2τ_(ε−up−1)m +9(Δf/f)T_(FM)m)T_(FM) 2(Δf/f)T_(FM) ² m . . . and so on

[0181] Going back to equation (16), and setting these errors to zero,the deswept signal can be written,

deswept r(t)=H(ω′)e ^(jφ(t)+j). constant phase term  (19)

[0182] Therefore, the observed deswept signal (with other errorsremoved) now consists of the time analog of the transfer function H(jω′)at some unknown angle, (the constant phase term), along with a phasenoise component φ(t).

[0183] Without loss of generality, the term φ(t) may be taken as thedifference between the phase noise at the beginning of the sweep and thephase noise at some point in time t into the sweep since the constantphase term incorporates the starting phase. Then using conventionaltechniques ³⁸ ³⁹, the sweep duration T_(FM) can be chosen sufficientlysmall to make the accumulated phase noise as small as wanted. However,as Table 4 indicated, if the sweep is too short compared to the delayspread, another type of phase error is introduced, a phase twist in thecomputed impulse response according to equation (8). Hence there is arange over which T_(FM) can be traded off to minimize both of theseeffects.

[0184] Finally, the remaining constant phase term in equation (19) isnot important to the operation of this invention. Instead, it can beconsidered as a normally occurring phase error found in any coherentsystem of communication, that must be removed in some fashion, say by aphase locked loop.

[0185] The analysis in this section showed that it is possible to removemisalignment errors that cause artifacts with a minimum of three sweeps,two up and one down (or two down and one up) during the tracking stage.The sweeps do not necessarily have to be contiguous but could occur inmany different combinations, a few of which are shown in FIG. 14, aslong as appropriate care is taken to account for the accumulation oftiming errors between sweeps.

[0186] 4. Description of a Prototype VNA

[0187] The inventors have developed a prototype working model of the VNAdescribed in this Application and obtained permission in 2001, in theform of a Special Temporary Authority from the FCC, to test it outdoors.See FIG. 15. Note, FIG. 15 is not in the form of an applique to acommunication system. Its purpose is to test algorithms for artifactremoval.

[0188] The Direct Digital Synthesizer, FIG. 15, is made by AnalogDevices, and is their Model Number AD9852/PCB. The 30 dB amplifier with0.2 watts output is made by Q-Bit and is their model number QB-500-2 RFAmplifier 8718-11. The transmit antenna is a 300 ohm television rabbitears. A balun is used to match the Q-Bit amplifier to the antenna. Thesame transmit unit was used to run all of the tests approved for 2001operation by the FCC. Frequencies requested were 82 MHz to 88 MHz. Thefrequency sweep waveform is completely settable by the Analog Devicesdirect digital synthesizer.

[0189] Maximum effective radiated power (ERP) or equivalentisotropically radiated power (EIRP) from the television rabbit ears,which is the equivalent of a short dipole with maximum gain 1.77 dB ⁴⁰,along with a radiated power of two tenths of a watt, results in an EIRPof −5.2 dBw or about 25 dBm. The FCC emission designator is for arepetitive FM energy dispersal function with linear frequency up-rampand down-ramp having continuous phase at the transitions. According toparagraph 2.201 of 47 CFR Chapter 1 (10-1-99 Edition), this signal hasemission designator F3N: F for frequency modulation, 3 for singlechannel containing analog information, and N for no informationtransmitted since no additional information is carried other than thesawtooth.

[0190] Examples of in-phase and quadrature waveforms recorded during atest with transmitter and receiver separated by 35 meters are shownnext. Sampled output of the baseband filter during an acquisitioncoincidence depicted schematically in FIG. 10(b), is shown here in FIG.16.

[0191] Duration of the sweep is roughly one second long and represents asweep covering 6 MHz. The prototype invention was first calibrated bymeasuring a known transfer function.

[0192]FIG. 17 shows the artificial coincidence in more detail,illustrating that the rotation direction changes at the precise time ofthe coincidence maximum. Sampled output of baseband filter duringtracking (step 6 of Table 3) is shown in FIG. 18. The heterodynedsignal's frequency is now totally within the baseband filter. See alsoFIGS. 8 and 11 for definition of the tracking condition. Also shown inFIG. 18 is visible parabolic phase versus time due to Δf/f . FIG. 19shows the logical processor results after parabolic phase distortion dueto Δf/f has been removed, and all residual spin removed. FIG. 19 is thecomputed H(jω) real part and imaginary part, corresponding to thein-phase and quadrature signal respectively.

[0193]FIG. 20 shows the in-phase and quadrature signals plotted in polarcoordinates. Note, significantly, changes in amplitude and phase overthe band. Some multipath signal cancellation is evident in the link. Atthe RF frequency used in the prototype, phase noise is negligible.

[0194] 5. Additional Embodiments

[0195] A person familiar with the design of signal processingelectronics will see that there are many implementations possible tothis invention. The implementation described in this Application ispreferred by the authors in the prototype version they have built, whereflexibility is necessary to allow hardware to be field-calibrated andsoftware to be tested and debugged. However, as the authors point out,many variations of the underlying invention can be configured to meetdifferent wireless system constraints and system needs.

[0196] 5.1. Changes in the Format of the Test Signal

[0197] The inventors have already noted that settability parameters onthe test signal and the facsimile signal produce a wide variety ofadditional embodiments that:

[0198] enable flexibility in the way signals get interlaced withcommunication, see FIG. 5(a),

[0199] enable flexibility between start-up of communications andcontinuation of communications,

[0200] enable flexibility to interlace signals with both forward-goingand backward-going communications over the same frequency band as shownin FIG. 5(b).

[0201] Also, the inventors pointed out that the use of linear FM sweepsshown in FIG. 2 and equation (1) were preferred embodiments but thatthere were many other suitable waveforms to use for the VNA probesignal. A simple example of an alternative embodiment is the use ofCDMA⁴¹ as the test signal. FIG. 21 shows a CDMA burst interlaced withcommunications. FIG. 22 shows the facsimile CDMA signal undergoingcontrol variations of start time, frequency offset and time base.

[0202] With CDMA, there would also be an acquisition mode and a trackingmode operationally having the same function and purpose as our preferredembodiment. The reason someone would choose to implement the inventionwith CDMA instead of linear FM is that they may already have CDMA chipsets and software which can more easily be adapted to the logicalprocessor control functions.

[0203] Other test signals and facsimile signals are possible, but allwould have to have:

[0204] settability parameters similar to that shown in FIGS. 3(a) and3(b),

[0205] control parameters on the facsimile similar to that shown in FIG.4,

[0206] operation by means of embedding into an active communicationschannel,

[0207] start-up functionality by means of acquisition and track modes inthe logical processor,

[0208] measurements collection and conveyance capability when trackingerrors are driven close to zero, indicating removal of artifacts.

[0209] One final word about the logical processor's determination thatartifact removal is complete. The way coherent digital communicationsreceivers work, they normally have to acquire and track out residualerrors in frequency, timing, and unknown phase. As equation (19) showed,a random constant phase is not considered the problem of the VNA but ofthe demodulator. Similarly, a small frequency offset or time offset canbe left for the demodulator to find and remove. The job of the VNA inthis invention is to find out how amplitude and phase shift vary acrossthe band of the communications, so any small fixed frequency offset andtime offset, which translate into a linear phase versus frequency termadded to the underlying phase versus frequency caused by multipath, areimmaterial to the final result. Note also, that time offset andfrequency offset, even if they are large, do not affect the amplitudeversus frequency of the measurement.

[0210] Stated in different terms, the answer produced by the VNA in thisinvention is not unique. While this fact may seem disconcerting atfirst, an experienced communications engineer familiar with the waymodems work will understand that non-uniqueness is not a liability inthe channel measurement as long as the modem sees a signal within itsnormal capture range. The inventors envision that the adaptivityfunction 9 in FIG. 1 in its control of the data modem 10, will performfrequency equalization using the H(jω) just measured, and will alsocorrect time and frequency offsets that were estimated in the logicalprocessor.

[0211] The statements above point out the synergism realized in the useof this invention. To take advantage of the adjustments in time andfrequency offset that were measured in the logical processor as itremoved artifacts, it is necessary to synchronize the communicationsstart time and frequency with the measurement signal time and frequency.Hence the precise start of the communication 56 and 57, or 62 and 63 inFIG. 5, must be synchronized with the test signal's time and frequency.This is possible since the transmit end adaptivity function 34 controlsparameters on the data encoder 20 and the data modulator 4 before thesignals arrive at the interlacing function 15. A person familiar withthe design of data modems, realizing that provision must be made fortiming and frequency offset at the receive end, would therefore know howto use the logical processor's instructions 33 to the adaptivityfunction 9 to enhance the design of the data modem receiver 10 in themanner discussed above.

[0212] 5.2. Alternate Embodiment: The New VNA as a Pure ChannelMeasurement Device

[0213] The inventors envision that there will be a need to run the newVNA design in a pure data-gathering mode. FIG. 23 shows an additionalembodiment of the invention with end-user network connections removedand the associated data multiplexing toggle switches removed. Note, thatcommunications components are here used entirely for internallygenerated control and measurements distribution. A storage device forthe measurements is not shown in the figure. Once again, a criticalelement in the invention's usefulness is the fact that the propagationpath is being measured between two internal points, 96 and 95 in FIG. 1and also in FIG. 24. This fact should not lessen its usefulness as apure

REFERENCES CITED IN SPECIFICATION

[0214]¹ ITU Recommendation ITU-R P.526-6, Propagation by Diffraction(Question ITU-R 202/3), 1978 to 1999.

[0215]² ITU Recommendation ITU-R P.530-8, Propagation Data andPrediction Methods Required for the Design of Terrestrial Line-Of-Sightsystems, (Question ITU-R 204/3), 1978 to 1999.

[0216]³ Lee, W. C. Y., Mobile Communications Engineering Theory andApplications Second Edition, McGraw Hill Telecommunications, Chapter 4,section 4.2.

[0217]⁴ Gibson, Jerry D., The Mobile Communications Handbook SecondEdition, CRC Press, IEEE Press, Chapter 18, Rayleigh Fading Channels byBernard Sklar, 1999.

[0218]⁵ ITU Recommendation ITU-R P.526-6, Propagation by Diffraction(Question ITU-R 202/3), 1978 to 1999.

[0219]⁶ Bertoni, Radio Propagation for Modern Wireless Systems, PrenticeHall PTR, Upper Saddle River, N.J., pp. 132-136.

[0220]⁷ ITU Recommendation ITU-R P.530-8, Propagation Data andPrediction Methods Required for the Design of Terrestrial Line-Of-Sightsystems, (Question ITU-R 204/3), 1978 to 1999.

[0221]⁸ See Gibson, op cit., Chapter 33 by Levesque, A. H., andPahlavan, K., Section 33.5, the Ricochet Network.

[0222]⁹ ITU-T Recommendation V.34, Series V: Data Communication Over theTelephone Network, (02/98), section 10.1.2.4 Line Probing Signals, and11.6 Rate Negotiation. See also CCITT V.32, CCITT V.22, CCITT V.21.

[0223]¹⁰ U.S. Pat. No. 5,347,539, High Speed Two Wire Modem, Sridhar, M.R., Mukherjee, A., Moran, J. L., Sep. 13, 1994.

[0224]¹¹ GTE Lenkurt, Engineering Considerations for MicrowaveCommunications Systems, 1972, GTE San Carlos Calif.

[0225]¹² Reudink, D. O., Properties of Mobile Radio Propagation Above400 MHz, IEEE Transactions on Vehicular Technology, November 1974,appearing in Rappaport, T. S., Cellular Radio and PersonalCommunications, A Book of Selected Readings, IEEE 1995.

[0226]¹³ IEEE Vehicular Technology Society Committee on RadioPropagation, Coverage Prediction for Mobile Radio systems Operating inthe 800/900 MHz frequency Range, IEEE transactions on VehicularTechnology, Vol. 37, No. 1, February 1988, reprinted in Cellular Radioand Personal Communications A Book of Selected Readings, edited byTheodore S. Rappaport, IEEE Press, 1995.

[0227]¹⁴ Greenstein, L. J., Michelson, D. G., and Erceg, V.,Moment-Method Estimation of the Ricean K-Factor, IEEE CommunicationsLetters, Vol. 3, No. 6, June 1999, p 175-176.

[0228]¹⁵ Schwartz, M., Bennett, W. R., and Stein, S., Communicationssystems and Techniques, Mc-Graw Hill Book Company, Inter-UniversityElectronics Series, 1966, Chapter 9, Fading Communications Media.

[0229]¹⁶ Erceg, V., et al, Channel Models for Fixed WirelessApplications (final IEEE 802.16 TG3 ad hoc version), IEEE802.16.3c-01/29r1, Feb. 23, 2001, available on the web site of NIST,Boulder Colorado.

[0230]¹⁷ Weinstein, S. B., and Ebert, P. M., Data Transmission byFrequency-Division Multiplexing Using the Discrete Fourier Transform,IEEE Transactions on Communications Technology, Vol. Com-19, No. 5,October 1971.

[0231]¹⁸ Van Nee, R., and Prasad, R., OFDM For Wireless MultimediaCommunications, Artech House Publishers, Boston, 2000, section 5.2.1Two-Dimensional Channel Estimators, and 5.2.3 Special Training Symbols.

[0232]¹⁹ Yang, B., Letaif, K. B, Cheng, R. S. and Cao, Z., TimingRecovery for OFDM Transmission, IEEE Journal on selected Areas incommunications, Vol. 18, No. 11, November 2000, pp. 2278-2291.

[0233]²⁰ Yeh, C-S, Lin, Y., Channel Estimation Using Pilot Tones in OFDMSystems, IEEE Transactions on Broadcasting, Vol. 45, No. 4, December1999. pp. 400-409.

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[0235]²² Simon, M. K., Hinedi, S. M., Lindsey, W. C., DigitalCommunication Techniques Signal Design and Detection, Prentice HallEngelwood Cliffs, N.J. 1995, section 9.4 Equalization Techniques.

[0236]²³ Proakis, J. G., Digital Communications Third Edition,McGraw-Hill Series in Electrical and Computer Engineering, 1995, Chapter11, Adaptive Equalization.

[0237]²⁴ Sklar, op cit.

[0238]²⁵ See for example the IEEE 802.16 working group discussion onreceived signal strength indication (RSSI) as an adaptivity approach.The URL for this group is.

[0239]²⁶ Agilent Technologies (Formerly Hewlett Packard) 2001 Catalog,pages 266-302.

[0240]²⁷ Pahlavan and Levesque Wireless Communication Networks JohnWiley and Sons 1995.

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[0242]²⁹ Xu, Hao, Terrestrial Radio Wave Propagation at millimeter-waveFrequencies, Department of Electrical Engineering and ComputerEngineering, Virginia Tech ETD,

[0243] http://scholar.lib.vt.edu/theses/available/etd-05042000-16180036/

[0244]³⁰ U.S. patents using the words network analyzer (sample) incommunications and radar patents:

[0245] (a) High-Speed Broadband Wireless Communication SystemArchitecture, #6240274, Izadpanah.

[0246] (b) Swept-Step Radar System and Detection Method using the Same,#6225941, Gogineni.

[0247] (c) Practical Space Time Radio Method for CDMA CommunicationCapacity Enhancement, #6108565, Scherzer.

[0248] (d) Radio Environment Analysis Apparatus, #6084928, Kuwahara.

[0249] (e) Network Analyzer Measurement Method Using Adaptive SignalProcessing, #6065137, Dunsmore.

[0250]³¹ Proakis, J. G., Digital Communications Third Edition,McGraw-Hill Series in Electrical and Computer Engineering, 1995, page586, Optimum receiver for an AWGN Channel with ISI.

[0251]³² Viterbi, Andrew J., and Omura, Jim K., Principles of DigitalCommunication and Coding, McGraw Hill, New York, 1979, Section 4.4.

[0252]³³ Schlegel, Christian, Trellis Coding, IEEE Press, © 1997, IEEE,New York City.

[0253]³⁴ Ziemer, Rodger E., and Tranter, William H., Principles ofCommunications, Systems Modulation and Noise, John Wiley and Sons, ©2002, Section 10.3.12 page 558.

[0254]³⁵ Helstrom, C. W., Statistical Theory of Signal Detection,Pergamon Press, Oxford, 1968, page 357.

[0255]³⁶ Skolnik, Introduction to Radar Systems Second Edition, page422, McGraw Hill, 1980.

[0256]³⁷ All signals described in this patent Application are shown incomplex envelope bandpass equivalent notation. See for example,Oppenheim, A. V., and Schafer, R. W., Digital Signal Processing,Prentice-Hall, Engelwood Cliffs, N.J., 1975, page 363.

[0257]³⁸ Gardner, Floyd, Phaselock Techniques, Wiley-IntersciencePublication, John Wiley and Sons, 1979, page 100, section Heading:Oscillator Phase Noise.

[0258]³⁹ Wolejsza, Chester A., Effects of Oscillator Phase Noise on PSKDemodulation, Comsat Technical Review Volume 6 Number 1, Spring 1976,page 107-125.

[0259]⁴⁰ Gagliardi, Robert M., Satellite Communications, Second Edition,Van Nostrand Reinhold, New York, © 1991, Table 3.1 Antenna Patterns.

[0260]⁴¹ Sklar, Bernard, Digital Communications Fundamentals andApplication, Second Edition, © 2001, Prentice Hall PTR, Upper SaddleRiver, N.J., Section 11.1.5. page 672.

1. A vector network analyzer which measures amplitude and phase-shiftversus frequency of a radio propagation link typical of licensed-band orunlicensed-band wireless communications, said analyzer being used in anapplique mode appended to a wireless radio frequency communicationslink, and comprising: a test signal injector which interlaces radiofrequency testing signals with communications signals, a test signalreceiver which receives said interlaced test signals and formsmeasurements of the amplitude versus frequency and phase-shift versusfrequency characteristics of the propagation path, a measurementsconveyance part which sends extracts of said measurement to operationalcontrol points.
 2. A vector network analyzer described in claim 1 thathas a test signal receiver part consisting of: an adjustable signalgenerator which generates a facsimile of said transmitted RF testingsignal and which has adjustable parameters: start time, frequencyoffset, and time base, a downconverter at said receiving end which usessaid adjustable facsimile signal as the heterodyning signal, a pluralityof analog-to-digital converters which samples in-phase and quadraturecomponents of said downconverted signal for analysis in a logicalprocessor, for which a personal computer is one example, a controllerfrom said logical processor to said adjustable facsimile signalgenerator which changes the time offset, frequency offset or time baseparameters of said facsimile signal generator.
 3. An injector of radiofrequency test signals along with its associated facsimile generator atsaid receiving end in claim 1 for which: the time duration of theinjected and facsimile test signals is settable, the frequency occupancyof the injected and facsimile test signals is settable, the timeseparation between one injected test signal and the next injected testsignal is settable, the time separation between one facsimile testsignal and the next facsimile test signal is also settable to be eitherthe same or different from said time separation of said injected testsignal time separation, the rate of injection of test signals and rateof generation of facsimile signals are both settable, the transmit powerlevel of the test signal is settable independent of the transmit powerlevel of the communications, settability parameters are entered into thetest signal generator and facsimile generator either before the start ofuse, or are changed at some intermediate time during subsequent use. 4.A test signal receiver logical processor of claim 2 which operates onthe analog-to-digital sampled in-phase and quadrature heterodyned signalto perform operations of the following type: estimation operations whoseoutcome is an estimate of the frequency difference between thetransmitted test signal and the facsimile test signal, estimationoperations whose outcome is an estimate of the time shift between thetransmitted test signal and the facsimile test signal, estimationoperations whose outcome is an estimate of the time base differencebetween the transmitted test signal and the facsimile test signal. 5.Said logical processor making a plurality of adjustments to saidfacsimile signal generator, which processing steps are divisible intotwo categories: an acquisition processing category in which thefacsimile signal generator's time base settability parameter and starttime settability parameter are adjusted, a tracking processing categoryin which both start time and frequency offset settability parameters arevaried.
 6. A measurements conveyance part in claim 1 which sendsextracts of the measurement in numerical form produced by said logicalprocessor to an adaptivity function at the receiving end of the linkthat carries out one or more of the following operations with thenumerical data files so obtained: a recording operation that makes saidnumerical data available in a data base that relates said recordednumerical files to the parameters that were used by the test signalgenerator and the facsimile generator up to and including the time whenthe extracted numerical files were obtained from the logical processor,a control operation on the communications data demodulator at the testsignal receiving end of the link, said operation comprising the settingof a channel equalizer, a control operation on the communications datademodulator at the receiving end of the link, said operation comprisingthe setting of gain control, a data multiplexing operation in which saidextracted numerical files are multiplexed with an end-user's networkinterface data for transport across the radio frequency propagation pathultimately to a data demodulator at the transmitting end of the linkwhere said multiplexed data is demultiplexed into a plurality of datastreams, one of which contains the extracted numerical files, which isconveyed to a second adaptivity function, this additional adaptivityfunction being at the transmit end of the link, a control operation onthe data encoder at the receiving end of the link, said encoder takingmultiplexed data streams comprising said user's network interface datawith said extracted numerical files, said control operation typicallyinvolving the queuing of data and segmentation of data for subsequentmodulation, up-conversion and radio frequency transmission, a controloperation on the data modulator at the receiving end of the link, saidmodulator function typically consisting of mapping of encoded data intochannel waveforms for subsequent up-conversion and radio frequencytransmission, said control operation involving but not limited toselection of modulation alphabet, symbol duration, and mapping rules. 7.A measurements conveyance part in claim 1 which has a second adaptivityfunction at the transmitting end of the link which receives saiddemultiplexed numerical file information and which carries out one ormore of the following distributive operations with the numerical datafiles so obtained, these all occuring at the transmitting end of thelink: a recording operation that makes the numerical data available in adata base that relates said recorded numerical files to the parametersthat were used by the test signal generator and the facsimile generatorup to and including the time when the extracted numerical files wereobtained from the logical processor, a control operation on the datademodulator at the transmitting end of the link, said operationcomprising the setting of a channel equalizer, a control operation onthe data demodulator at the transmitting end of the link, said operationcomprising the setting of gain control, a data multiplexing operation inwhich said settability parameters are multiplexed with an end-user'snetwork interface data for transport over the radio frequency linkultimately to a data demodulator at the receiving end where saidmultiplexed data is demultiplexed into a plurality of data streams, oneof which contains the settability parameters to be used by the facsimilesignal generator, a control operation on the data encoder at thetransmitting end of the link, said encoder taking multiplexed datastreams comprising said user's network interface data with saidsettability parameters, said control operation typically involving thequeuing of data and segmentation of data for subsequent modulation,up-conversion and radio frequency transmission, a control operation onthe data modulator at the transmitting end of the link, said modulatorfunction typically consisting of mapping of encoded data into channelwaveforms for subsequent up-conversion and radio frequency transmission,said control operation involving but not limited to selection ofmodulation alphabet, symbol duration, and mapping rules.
 8. Ameasurements conveyance part in claim 1 which contains radio frequencydevices for accessing a band of frequencies, said devices enabling thetime sharing of the band of frequencies between transmissions of saidtest signals interlaced with communications on the one hand withconveyance of said multiplexed numerical files representing channelmeasurements with end-user network interface data on the other hand,said accessing devices comprising: radio frequency toggle switches atboth ends of the link which operate in a time synchronized fashion,wireless radio frequency antennas with appropriate pass-band filters andstop-band filters needed if necessary for purposes of eliminating orreducing interference into and from adjacent bands, high poweramplifiers and low noise amplifiers.
 9. Interpretive means foradaptively controlling queuing and segmentation of both forward-goingand backward-going communications based in part on measurement data innumerical files produced in said logical processor.
 10. A preferredembodiment of the invention of claim 1 in which injected radio frequencytest signal and associated facsimile signal are composite collections ofupward-going linear FM sweeps and downward-going linear FM sweeps. 11.Said logical processor for said preferred embodiment test signals whichperforms the following operations: finding an estimate of saidheterodyne signal's center frequency, finding an estimate of saidheterodyne signal's start time, finding an estimate of said heterodynesignal's center frequency rate of advancement or retardation.
 12. A testsignal receiver logical processor which operates on theanalog-to-digital sampled in-phase and quadrature heterodyned signalwhen the test signals are chosen to be of the preferred embodiment type,upward-going linear FM sweeps and downward going linear FM sweeps, saidparticular operations being further illustrated by a preferredembodiment set of processing steps: counting revolutions of the vectorrepresenting samples of said in-phase and quadrature numbers, therevolutions being taken about an origin point in a two-dimensional planerepresenting in-phase and quadrature samples, said revolution countbeing an approximation to the frequency offset of the heterodynedsignal, separating revolution counts according to whether they tookplace during a particular up-going FM sweep or down-going FM sweep,separating the aforementioned revolution counts according to thesequence in which the files were recorded by said analog-to-digitalconverter.
 13. An alternative embodiment vector network analyzer whichfinds amplitude and phase-shift versus frequency of a radio propagationlink typical of licensed-band or unlicensed-band wirelesscommunications, said analyzer comprising: a test signal injector whichinterlaces radio frequency testing signals with control signals, a testsignal receiver which receives said interlaced test signals and formsmeasurements of the amplitude versus frequency and phase-shift versusfrequency characteristics of the propagation path, a conveyance partwhich sends control signals and extracts of the measurements in the formof numerical files to operational control points.
 14. A vector networkanalyzer described in claim 13 that has a test signal receiver partconsisting of: an adjustable signal generator which generates afacsimile of said transmitted RF testing signal and which has adjustableparameters: start time, frequency offset, and time base, a downconverterat said receiving end which uses said adjustable facsimile signal as theheterodyning signal, a plurality of analog-to-digital converters whichsamples in-phase and quadrature components of said downconverted signalfor analysis in a logical processor, for which a personal computer isone example, a controller from said logical processor to said adjustablefacsimile signal generator which changes the time offset, frequencyoffset or time base parameters of said facsimile signal generator. 15.An injector of radio frequency test signals along with its associatedfacsimile generator at said receiving end in claim 13 for which: thetime duration of the injected and facsimile test signals is settable,the frequency occupancy of the injected and facsimile test signals issettable, the time separation between one injected test signal and thenext injected test signal is settable, the time separation between onefacsimile test signal and the next facsimile test signal is alsosettable to be either the same or different from said time separation ofsaid injected test signal time separation, the rate of injection of testsignals and rate of generation of facsimile signals are both settable,the transmit power of the test signal is settable, settabilityparameters are entered into the test signal generator and facsimilegenerator either before the start of use, or are changed at someintermediate time during subsequent use.
 16. A test signal receiverlogical processor of claim 14 which operates on the analog-to-digitalsampled in-phase and quadrature heterodyned signal to perform operationsof the following type: estimation operations whose outcome is anestimate of the frequency difference between the transmitted test signaland the facsimile test signal, estimation operations whose outcome is anestimate of the time shift between the transmitted test signal and thefacsimile test signal, estimation operations whose outcome is anestimate of the time base difference between the transmitted test signaland the facsimile test signal.
 17. Said logical processor of claim 14making a plurality of adjustments to said facsimile signal generator,which processing steps are divisible into two categories: an acquisitionprocessing category in which the facsimile signal generator's time basesettability parameter and start time settability parameter are adjusted,a tracking processing category in which both start time and frequencyoffset settability parameters are varied.
 18. A measurements conveyancepart in claim 13 which sends extracts of the measurement in numericalform produced by said logical processor of claim 14 to an adaptivityfunction at the receiving end of the link that carries out one or moreof the following operations with the numerical data files so obtained: arecording operation that makes said numerical data available in a database that relates said recorded numerical files to the parameters thatwere used by the test signal generator and the facsimile generator up toand including the time when the extracted numerical files were obtainedfrom the logical processor, a control operation on the data demodulatorat the receiving end of the link, said operation comprising the settingof a channel equalizer, a control operation on the data demodulator atthe receiving end of the link, said operation comprising the setting ofgain control, a data conveyance operation in which said extractednumerical files are conveyed over the radio frequency channel ultimatelyto a second adaptivity function at the transmitting end of the link, acontrol operation on the data encoder at the receiving end of the linkinvolving the queuing of numerical file data of channel measurements,and segmentation of said numerical data for subsequent modulation,up-conversion and radio frequency transmission, a control operation onthe data modulator at the receiving end of the link, said modulatorfunction typically consisting of mapping of encoded data into channelwaveforms for subsequent up-conversion and radio frequency transmission,said control operation involving but not limited to selection ofmodulation alphabet, symbol duration, and mapping rules.
 19. Ameasurements conveyance part in claim 13 which has a second adaptivityfunction at the transmitting end which receives said numerical fileinformation of channel measurements, which carries out one or more ofthe following distributive operations with the numerical data files soobtained, these all occuring at the transmitting end of the link: arecording operation that makes the numerical data available in a database that relates said recorded numerical files to the parameters thatwere used by the test signal generator and the facsimile generator up toand including the time when the extracted numerical files were obtainedfrom the logical processor, a control operation on the demodulator atthe transmitting end of the link which receives numerical files ofchannel measurements, said operation comprising the setting of a channelequalizer, a control operation on the demodulator at the transmittingend of the link which receives numerical files of channel measurements,said operation comprising the setting of gain control, an operation inwhich said settability parameters of claim 15 are queued for transportover the radio frequency link ultimately to a data demodulator at thereceiving end where said settability parameters are conveyed to thefacsimile signal generator, a control operation on the data encoder atthe transmitting end of the link, said encoder taking data streamscomprising settability parameters, said control operation typicallyinvolving the queuing of data and segmentation of data for subsequentmodulation, up-conversion and radio frequency transmission, a controloperation on the data modulator at the transmitting end of the link,said modulator function typically consisting of mapping of encoded datainto channel waveforms for subsequent up-conversion and radio frequencytransmission, said control operation involving but not limited toselection of modulation alphabet, symbol duration, and mapping rules.20. A measurements conveyance part in claim 13 which contains radiofrequency devices for accessing a band of frequencies, said devicesenabling the time sharing of the band of frequencies betweentransmissions of said test signals interlaced with settability controlsignals on the one hand with conveyance of said numerical filesrepresenting channel measurements on the other hand, said accessingdevices comprising, radio frequency toggle switches at both ends of thelink which operate in a time synchronized fashion, wireless radiofrequency antennas with appropriate pass-band filters and stop-bandfilters needed if necessary for purposes of eliminating or reducinginterference into and from adjacent bands, high power amplifiers and lownoise amplifiers.
 21. A preferred embodiment of claim 13 in whichinjected radio frequency test signal and associated facsimile signal arecomposite collections of upward-going linear FM sweeps anddownward-going linear FM sweeps.
 22. Said logical processor for saidpreferred embodiment test signals in claim 21 which performs thefollowing operations: finding an estimate of said heterodyne signal'scenter frequency, finding an estimate of said heterodyne signal's starttime, finding an estimate of said heterodyne signal's center frequencyrate of advancement or retardation.
 23. A test signal receiver logicalprocessor for the preferred embodiment of claim 21 which operates on theanalog-to-digital sampled in-phase and quadrature heterodyned signalwhen the test signals are chosen to be combinations of upward-goinglinear FM sweeps and downward going linear FM sweeps, said particularoperations being further illustrated by a preferred embodiment set ofprocessing steps: counting revolutions of the vector representingsamples of said in-phase and quadrature numbers, the revolutions beingtaken about an origin point in a two-dimensional plane representingin-phase and quadrature samples, said revolution count being anapproximation to the frequency offset of the heterodyned signal,separating revolution counts according to whether they took place duringa particular up-going FM sweep or down-going FM sweep, separating theaforementioned revolution counts according to the sequence in which thefiles were recorded by said analog-to-digital converter.
 24. A vectornetwork analyzer of claim 1 used as an applique to a communications linkwherein the propagation characteristics are typical of unlicensed band,licensed band, and additionally mobile communication bands of operation.25. The vector network analyzer of claim 24 suitable for mobilepropagation channels, which also has said settability of the rate ofoccurrence parameter permitting said vector network analyzer to adjustthe speed of interlacing of test signals to enable it to characterizepropagation conditions in a mobile channel, such channels generallychanging at a more rapid pace than non-mobile channels.